Television receivers and methods for processing signal sample streams synchronously with line/frame patterns

ABSTRACT

Television receivers receieve a television signal and downconvert the television signal to generate a complex baseband signal. The complex baseband signal is differeniated and then integrated to reduce a direct current (DC) offset in the complex baseband signal. The television receiever may include a periodic known information field. The known information field is correlated with the integrated signal to determine a residual offset therein that is associated with a constant used in integrating the differentiated complex baseband signal. The residual offset is substracted from this constant when integrating the differeniated complex baseband signal.

BACKGROUND OF THE INVENTION

[0001] The present invention relates generally to the field oftelevision receivers, and, more particularly, television receivers andmethods for reducing ghosting and for mitigating the deleterious effectson television signal reception due to multipath propagation.

[0002] In the United States (U.S.), there are numerous televisionstations that transmit video modulated signals from towers distributedabout the country. In many locations, it may be difficult or impossibleto use a fixed-pointing directional antenna, as the televisiontransmission towers may lie in various directions. In general, there isno reason to expect better angular correlation of sites with theintroduction of digital or HDTV standards.

[0003] Antenna rotators are often used to overcome this difficulty, butthe rotators typically operate slower than the speed at which a viewermay wish to “channel-surf.” The need to re-orient an antenna to receivedifferent channels may be an inconvenience and may present otherdifficulties when an antenna is shared between many television sets indifferent rooms or dwellings, such as an apartment building where manydifferent viewers will typically be tuned to many different televisionchannels simultaneously.

[0004] In apartment buildings without communal antenna systems, a viewermay need to resort to an indoor antenna to receive broadcast televisionsignals, which may then be subject to distortion due to reflection fromnearby objects in the same room or even on the other side ofradio-transparent walls. In locations obscured by surrounding highterrain, it may be difficult or impossible to receive a directtelevision signal wave with any reasonable external antenna mast height,although television signals may be received by diffraction over a hillor by reflection. These diffracted and/or reflected signals may beimpaired due to multipath propagation, which may result in attenuationof some frequency components of the video signal and/or ghosting if themultipath delays are a non-negligible fraction of the line scan period.Often, the preference for a high-gain, outdoor, directional antennastems not from the need for greater signal strength, but from the needto exclude signal reflections by angle-of-arrival discrimination toreduce ghosting.

[0005] U.S. Pat. No. 5,119,196 to Ayanoglu et al. (hereinafter '196patent), the disclosure of which is hereby incorporated herein byreference, describes finite impulse response (FIR) and infinite impulseresponse (IIR) equalization techniques for television picture ghostcancellation. The '196 patent describes limitations of an IIR channelinverse equalizer when the poles of the channel-descriptive z-polynomialapproach the unit circle. The '196 patent also describes factorizing thechannel-descriptive polynomial into a causal factor having poles insidethe unit circle and an anti-causal factor having poles outside the unitcircle, and processing a scan line of signal samples in time-reversedorder to implement the anti-causal part. The '196 patent explains thatthe limitations of these conventional equalizers may be partiallyeliminated by assuming that the line sync pulse is a zero video signalfor a period at each end of the scan line that can be regarded as aguard time between lines. Thus, for ghost delay spread shorter than thisguard time, signal equalization may be performed on a line-by-line basiswith no carryover from or to adjacent lines. This technique equalizesthe video information in the line, but may not remove interference fromthe sync pulse. Unfortunately, according to the U.S. NTSC standard, thesync pulse is only a zero signal only for a period after detection andif the sync pulse is clamped by a DC restoration circuit such that itspeak value is zero. It may be desirable to equalize a television signalpredetection at which time the sync pulse is not a zero signal, butinstead is a period of maximum transmission.

[0006] The techniques described in the '196 patent may also be limitedin the delay that they can equalize. The '196 patent describes aprocessing-intensive technique for adapting equalizer weights for aleast-squares reproduction of a test signal. The '196 patent suggestscomplementing the ghost reduction equalizers with an adaptive antenna,but does not provide implementation details for the adaptive antenna orsuggest that the adaptive antenna may be a diversity antenna having twoseparate output signals. The '196 patent appears to suggest processingthe television video signal after detection, which may not allowexploitation of the relative phase difference between delayed signalpaths. Finally, the equalization system described in the '196 patent isdesigned to cancel echos by subtracting them away from the desiredsignal. Unfortunately, this approach may risk degradation of the desiredsignal in the echo cancellation process.

[0007] Other patents that discuss ghost removal include: U.S. Pat. No.5,253,063 to Ebihara et al. (hereinafter '063 patent), U.S. Pat. No.5,812,217 to Cahill, III (hereinafter '217 patent), and U.S. Pat. No.5,331,416 to Patel et al. (hereinafter '416 patent). Patents thatdiscuss antenna selection diversity include: U.S. Pat. No. 5,335,010 toLendemeier et al. (hereinafter '010 patent) and U.S. Pat. No. 5,818,543to Lee (hereinafter '543 patent). The '063 patent, '217 patent, '416patent, '010 patent, and '543 patent are hereby incorporated herein byreference.

[0008] Unfortunately, antenna selection diversity may be hard toreconcile with ghost equalization due to the changing ghostcharacteristics caused by antenna switching. Accordingly, there is aneed for improved television receivers, which may reconcile diversityreception with ghost removal in an economically efficient manner.

SUMMARY OF THE INVENTION

[0009] Embodiments of the present invention provide television receiversand methods of operating same in which a television signal is receivedand downconverted to generate a complex baseband signal. The complexbaseband signal is differentiated and then re-integrated to reduce adirect current (DC) offset in the complex baseband signal. Thetelevision receiver may comprise a periodic known information field. Theknown information field is correlated with the integrated signal todetermine a residual offset therein that is associated with a arbitraryconstant arising from re-integrating the differentiated complex basebandsignal. The residual offset is subtracted from re-integrated,differentiated complex baseband signal constant when integrating thedifferentiated complex baseband signal.

BRIEF DESCRIPTION OF THE DRAWINGS

[0010] Other features of the present invention will be more readilyunderstood from the following detailed description of specificembodiments thereof when read in conjunction with the accompanyingdrawings, in which:

[0011]FIG. 1 illustrates an example of multipath television reception;

[0012]FIG. 2 illustrates an example of diversity television receptionfor processing multipath signals;

[0013]FIG. 3 illustrates the frequency spectrum of an NTSC televisionsignal;

[0014]FIG. 4 is a block diagram that illustrates a homodyne televisionreceiver in accordance with embodiments of the present invention;

[0015]FIG. 5 illustrates a sync pulse of an NTSC television signal;

[0016]FIG. 6 illustrates the presence of a half-color subcarrier signalin the sync pulse;

[0017]FIG. 7 is a block diagram that illustrates a homodyne televisionreceiver in accordance with further embodiments of the presentinvention; and

[0018]FIG. 8 is a block diagram that illustrates a diversity homodynetelevision receiver in accordance with embodiments of the presentinvention.

DETAILED DESCRIPTION OF PREFERRED EMBODIMENTS

[0019] While the invention is susceptible to various modifications andalternative forms, specific embodiments thereof are shown by way ofexample in the drawings and will herein be described in detail. Itshould be understood, however, that there is no intent to limit theinvention to the particular forms disclosed, but on the contrary, theinvention is to cover all modifications, equivalents, and alternativesfalling within the spirit and scope of the invention as defined by theclaims. Like reference numbers signify like elements throughout thedescription of the figures. It will also be understood that when anelement is referred to as being “connected” or “coupled” to anotherelement, it can be directly connected or coupled to the other element orintervening elements may also be present. In contrast, when an elementis referred to as being “directly connected” or “directly coupled” toanother element, there are no intervening elements present.

[0020] Referring now to FIG. 1, an example of multipath televisionreception is illustrated. Typically, UHF signals are used in cellularwireless telephone systems or television broadcast systems. By itsnature, a broadcast system illuminates a wide area from an antenna site,and reflections from objects such as buildings or terrain features maybe received at the television receiver as well as the direct wave. Asshown in FIG. 1, a television 12 receives a direct wave signal from atransmitter 14 along with a second signal that is reflected from aterrain feature and a third signal that is reflected from a building 16.The superposition of delayed waves may cause distortion to theinformation modulation. In digital cellular systems, various equalizermethods have been developed to reduce degradation of digital data errorrates due to multipath distortion. In television broadcast systems,multipath propagation has long been recognized as the cause of picturedegradation known as ghosting. One approach to reducing ghosting is touse a directional antenna for television signal reception and to orientthe antenna to receive the strongest wave and to exclude other wavesarriving from different directions. This approach, however, discardsenergy arriving at the television receiver that could, in principle, beused to improve reception if it could be properly integrated.

[0021]FIG. 2 illustrates an example of diversity television receptionfor processing multipath signals. A television 22 comprises a firstreceiver 24 and a second receiver 26. The first receiver 24 receives adirect wave signal and a reflected signal from a terrain feature, whichare transmitted by a transmitter 28, through a first antenna 32. Thesecond receiver 26 receives a signal from the transmitter 28, which hasbeen reflected from a building 34, through a second antenna 36. Thesignals received through the first receiver 24 are delayed by a delaycircuit 38, and then the signals received through both the first andsecond receivers 24 and 26 are combined by the adder circuit 42.

[0022] Television receivers, in accordance with embodiments of thepresent invention, are based on a homodyne or direct conversion receiverarchitecture. Improvements to these architectures are described in thefollowing patents, which are hereby incorporated herein by reference:U.S. Pat. No. 5,241,702 to Dent (hereinafter '702 patent), U.S. Pat. No.5,568,520 to Lindquist et al. (hereinafter '520 patent), U.S. Pat. No.5,712,637 to Lindquist et al. (hereinafter '637 patent), U.S. Pat. No.5,749,051 to Dent (hereinafter '051 patent), and U.S. Pat. No. 5,918,169to Dent (hereinafter '169 patent).

[0023] The '702 patent describes a solution for a direct current (DC)offset problem, which may occur in homodyne or direct conversionreceivers. The homodyne DC offset problem refers to the directconversion of small received signals in the microvolt to millivolt rangeto complex baseband signals centered around zero frequency (DC) but at aDC offset on the order of several millivolts to hundreds of millivolts.Unfortunately, the large magnitude of the DC offset may drown out thedesired signal. For example, if the output signals of the homodynedownconvertor are to be digitized using an analog-to-digital (A/D)convertor, then the A/D full-scale setting preferably encompasses the DCoffset to avoid clipping. In this case, however, the desired signal mayonly occupy a few least significant bits. If sufficiently high dynamicrange A/D convertors can be used so that the number of least significantbits occupied by the desired signal is still adequate, then the DCoffset may be preserved through digital domain processing where it maybe distinguished from the desired signal components.

[0024] Another approach for removing homodyne DC offset involvesdifferentiating the homodyne downconvertor output signals to remove DCoffset, performing A/D conversion of the differentiated signals toobtained digitized samples of the differentiated signals, and thenintegrating the digitized samples using digital accumulation to undo thedifferentiation and restore the signal waveform. The restored signalwaveforms after digital accumulation have a DC offset that is equal toan arbitrary constant of reintegration, which is related to the value ofthe received signal at the time the accumulator was last zero or set tozero. This DC offset error is only as large as the signal value at thelast accumulator reset, and, thus, does not add significantly to the A/Dconvertor dynamic range requirements. This residual error may beestimated and subtracted out by using some feature of the desired signalthat allows it to be distinguished from the error.

[0025] Various examples of reception of digitally modulated data signalsand analog frequency modulated (FM) signals are described in the '702patent. In the case of digital signals, a known symbol pattern orsyncword may be included in transmissions and compared to the receivedwaveform after digital accumulation to determine multipath channelpropagation coefficients as well as DC offset. In the case of an analogFM signal, the complex signal values generated in a homodyne receivershould lie on a circle of constant amplitude and should vary only inphase angle. By identifying a locus of successive values, thedisplacement of the center of the locus circle from the origin may beidentified, which corresponds to the DC offset in the real and imaginary(I, Q) channels.

[0026] For purposes of illustration, embodiments of the presentinvention are described herein in the context of receiving a U.S. NTSCsignal, which uses vestigial sideband amplitude modulation. Thefrequency spectrum of an NTSC signal is illustrated in FIG. 3. It willbe understood, however, that embodiments of the present invention mayalso be used to receive analog television signals based on otherstandards, such as the European PAL standard and/or the French SECAMstandard. These standards are similar to the U. S. NTSC standard, withadjustments to the bandwidth, sound, and color subcarrier positions. Itwill be further understood that embodiments of the present invention mayalso be used to receive digital television signals. One digitaltelevision standard uses 8-level amplitude modulation. Accordingly,television receivers, in accordance with embodiments of the presentinvention, may be used to provide economic, multi-standard televisionreception.

[0027] Referring now to FIG. 4, a homodyne receiver 52, in accordancewith embodiments of the present invention, comprises a first antennainput and a second antenna input, which are coupled to a first inputfilter 54 and a second input filter 56, respectively. The receiver 52further comprises a digital frequency synthesizer 58 that, along withthe first and second input filters 54 and 56, is coupled to a quadraturedownconvertor circuit 62. The quadrature downconvertor circuit 62 iscoupled to a multi-mode digital signal processing (DSP) circuit 64 by abank of A/D convertors 66. A bank of digital-to-analog (D/A) convertors68 is coupled to the multi-mode DSP circuit 64.

[0028] The downconvertor circuit 62 comprises a pair of low noiseamplifiers (LNAs) 72 and 74 that are coupled to the first and secondinput filters 54 and 56, respectively. The output of the first LNA 72 iscoupled to the inputs of two mixer circuits 76 and 78. The output of thesecond LNA 74 is coupled to the inputs of two mixer circuits 82 and 84.The downconvertor circuit 62 further comprises a quadrature voltagecontrolled oscillator (QVCO) 86 (i.e., local oscillator), whichgenerates a first output signal corresponding to a sine wave centered ona desired frequency, such as a video carrier frequency, and a secondoutput signal corresponding to a cosine wave centered on the desiredfrequency. The sine wave signal output from the QVCO 86 is provided tomixer circuits 78 and 82 while the cosine wave signal output from theQVCO 86 is provided to mixer circuits 76 and 84. In this manner, an Isignal or in-phase signal is generated at the output of each of themixer circuits 76 and 84 while a Q-signal or quadrature signal isgenerated at the output of each of the mixer circuits 78 and 82.

[0029] The output signals from the mixer circuits 76, 78, 82, and 84contain frequency components corresponding to both the sum of thefrequencies of the input signals thereto and the difference of thefrequencies of the input signals thereto. Accordingly, low pass filters87, 88, 92, and 94 are coupled to the mixer circuits 76, 78, 82, and 84,respectively, to suppress those frequency components corresponding tothe sum of the frequencies and to pass those frequency componentscorresponding to the difference of the frequencies. Each I, Q signalpair output from the low pass filters 87, 88, 92, and 94 may be referredto as a complex baseband signal.

[0030] The A/D convertor bank 66 comprises four A/D convertors 96, 98,102, and 104, which generate digital samples from the output signalsfrom the low pass filters 87, 88, 92, and 94, respectively. These outputsamples are then processed by the multi-mode DSP circuit 64 to generate,for example, stereo sound signals, color signals, a line scan signal anda frame scan signal. The multi-mode DSP circuit 64 may also be used tocontrol the digital frequency synthesizer 58.

[0031] As discussed above, the digital frequency synthesizer 58 may beused to control the QVCO 86 such that the output signals from the QVCO86 are centered on the video carrier frequency used in conventionalanalog television operation and the first and second input filters 54and 56 may be configured to suppress interference on the sidebandfrequencies. In other embodiments of the present invention, the digitalfrequency synthesizer 58 may control the QVCO 86 such that the outputsignals from the QVCO 86 are centered in the middle of a televisionchannel and the low-pass filters 86, 88, 92, and 94 may be used toeffect symmetrical selectivity about the center of the channel. In thelatter case, each of the I and Q signals may be about 2.5-5 MHz wide,while in the former case, the I, Q signals may be about 6 MHz wide.Accordingly, these signals are sampled at the Nyquist rates of 6 MHz and12 MHz, respectively, or higher. A convenient rate that is greater thanthe minimum Nyquist rate and that may simplify later extraction of thecolor subcarrier information is an I, Q signal sample rate of 2 or 4times the color subcarrier rate, using a 2× or 4× color subcarrierfrequency crystal clock, which may be derived from the master crystalcoupled to the multi-mode DSP circuit 64. The color subcarrier frequencyin the NTSC standard is 3.579545 MHz, which results in sampling rates of7.15909 MHz or 14.31818 MHz. If the homodyne receiver is centered on thevideo carrier, then cross-coupled I, Q baseband filters may be used toimplement asymmetrical selectivity for suppressing lower adjacentchannel signals.

[0032]FIG. 5 illustrates a sync pulse used in an NTSC television signal.The sync pulse comprises a horizontal blanking pulse, which includes afront porch period, a horizontal sync pulse period, and a back porchperiod as shown. When a receiver is centered on the video carrierfrequency and frequency errors between the local oscillator and thevideo carrier frequency are removed by automatic frequency control (AFC)or phase locking, then a DC offset may appear on the video output signalto shift the video signal levels including the sync pulse levels. If thesync pulse has a peak level denoted by 100%, then the levels during thefront porch and back porch periods are at 75%. An additive DC offset Ioraises the sync pulse peak level to 1+Io and the front porch and backporch levels to 0.75+Io. The DC offset Io may, therefore, be deducedfrom the average value of four times the front/back porch level valueminus three times the sync pulse peak level value. The identified levelof Io may then be subtracted from the homodyne receiver output signal toreduce DC offset. An automatic gain control (AGC) circuit may also beused to average the corrected peak value of the sync pulse and to adjustthe receiver gain until it equals the expected or desired 100% videowaveform level.

[0033] Thus, according to embodiments of the present invention,processing I, Q signals synchronously with the NTSC sync pulse may allowestimation and subtraction of homodyne DC offset as will be described inmore detail below. This methodology for reducing DC offset may be usedindependent of the centering frequency used by the local oscillator.(e.g., QVCO 86 of FIG. 4). It will be further understood that theprinciples and concepts of the present invention may also be applied todigital television signals, which may be susceptible to multipathpropagation distortion, i.e., the same phenomenon that causes ghostingin analog television reception.

[0034] If the local oscillator is not centered on the video carrier,then frequency error exists between the local oscillator and the videocarrier. To reduce this frequency error, the I, Q signals output fromthe homodyne downconvertor circuit (e.g., downconvertor circuit 62 ofFIG. 4), may be rotated systematically to remove the frequency erroreither before or after being converted to digital samples. The frequencyerror may be detected by detecting phase rotation in successive (I, Q)values sampled at the sync pulse peaks. The error may be reduced bybringing the phase of the sync pulse peaks to the zero degree point onaverage. The homodyne DC offset may then, however, be shifted to thedifference frequency, which may cause the sync pulse levels to drift upand down in a repetitive sine wave fashion.

[0035] To reduce DC offset, the I, Q complex baseband signals outputfrom a downconvertor circuit may first be differentiated. Next, thedifferentiated signals may be processed by an A/D convertor bank toobtain a sequence of digital samples. After processing thedifferentiated and digitized signals to correct frequency error andmitigate ghosting, the resultant signal is a differentiated compositevideo signal from which the sound signal may be removed by a digitalnotch filter tune to reject the sound subcarrier frequency. Theresultant signal minus the sound signal may then be recombined withprecomputed differentiated sync pulse patterns to determine updatedtiming and frequency errors, and then integrated using digital valueaccumulation to restore the undifferentiated video signal.

[0036] The undifferentiated video signal now contains a DC offset,however, in the form of an arbitrary constant of integration. Thisconstant of integration is bounded, however, to a value on the order ofthe amplitude of the undifferentiated video signal. This residual DCoffset may be determined by averaging restored video signal samplestaken during the 8% of the line scan period H (see FIG. 5) that the syncpulse is at a peak value to determine a mean value L1 of the sync pulseamplitude, and also by averaging restored video signal samples takenduring the 2% of the line scan period H corresponding to the front porchperiod to obtain a mean value L2. The sync pulse should be at 75% of itspeak value during the front porch period. Accordingly, the DC offset tobe removed may be given by Equation 1 below:

DC offset=4*L 2−3*L 1  EQ.1

[0037] The DC offset value computed in Equation 1 may optionally beaveraged over many line scans or frames and then subtracted from therestored, undifferentiated video signal. After the residual DC offset isremoved, the mean level of the sync pulse amplitude may be used toadjust a gain control setting for the receiver chain until the peakamplitude of the sync pulse is equal to 100% of the desired video signalamplitude.

[0038] After the video signal has been processed to reduce DC offset andgain adjusted, the video signal may be further processed by, forexample, a multi-mode DSP circuit (e.g., the multi-mode DSP circuit 64of FIG. 4) to extract color information. The color information(chrominance) may be combined with the video signal (luminance) toproduce red, green, and blue (RGB) drive signals for a display. Thesedigital RGB signals may be processed by a D/A convertor bank (e.g., theD/A convertor bank 68 of FIG. 4) and optionally level shifted to drivethe electron guns of a cathode ray tube (CRT) or the analogous RGBintensity controls of other color displays, such as liquid crystaldisplays (LCDs). The digital processing may further include the removalof mono and/or stereo sound signals through the use of a notch filter.The digital sounds signals may also be converted into analog form usingone or more D/A convertors whose output signals may be amplified andused to drive loudspeakers.

[0039] In some embodiments of the present invention, televisionreceivers may have a local oscillator (e.g., QVCO 86 of FIG. 4) that iscentered at half the color subcarrier frequency above the video carrier.Low pass filters used to filter the I, Q signals (e.g., low pass filters87, 88, 92, and 94 of FIG. 4) may then be symmetrical and have notcheslocated at the channel spacing +/− (color subcarrier frequency)/2, whichmay improve the suppression of the upper and lower adjacent channelvideo carriers and their respective chrominance signals. Thesesymmetrical filters may be advantageous due to their simplicity. Asdiscussed above, the DC offset is shifted to the difference between thecenter frequency used by the local oscillator and the video carrier.Accordingly, the DC offset is shifted to 1.7897725 MHz, which is halfthe color subcarrier frequency. In addition, the I, Q signals may beprogressively phase-rotated to remove the half-color subcarrier offset.The video output samples may then be generated by combining the I and Qsamples to obtain a real video signal sampled at double the colorsubcarrier frequency.

[0040] Methods for reducing the residual DC offset due to the constantof integration when the local oscillator of a television receiver iscentered at half the color subcarrier frequency above the video carrier,in accordance with embodiments of the present invention, will now bedescribed.

[0041] When the I, Q signals are progressively phase rotated to removethe half-color subcarrier offset, the residual DC offset due to theconstant of reintegration may appear as an unwanted 1.7897725 MHzsignal. This may be difficult to distinguish from video signalcomponents around this frequency. Because video signal components arenot transmitted during the sync pulse interval, unwanted signalsoccurring during the sync pulse period may be determined and averagedover multiple pulses to determine an estimate related to the residual DCoffset. The estimated DC offset may then be subtracted from the I, Qsignals before progressive phase rotation to reduce the 1.7897725 MHzsignal component, which is present in the sync pulses.

[0042]FIG. 6 illustrates the presence of the half-color subcarriersignal in the sync pulse. This unwanted signal component may be detectedby subtracting digital samples one color subcarrier cycle apart, whichis half a cycle at the half-color subcarrier frequency. The colorsubcarrier burst at 3.58 MHz has equal values one cycle apart, whichcancel upon subtraction.

[0043] If I, Q samples at twice the color subcarrier frequency aredenoted by Z(i)={I(i), Q(i)}, then the following complex numbers may becomputed:

C=(Z(i)−Z(i+2)+Z(i+4)−Z(i+6)+Z(i+8)−Z(i+10)+Z(i+12)−Z(i+14))/8;

S=(Z(i+1)−Z(i+3)+Z(i+5)−Z(i+7)+Z(i+9)−Z(i+11)+Z(i+13)−Z(i+15))/8; and

P=(Z(i)+Z(i+1)+Z(i+2), . . . Z(i+15)/16.

[0044] where i is the first sample after the beginning flank of thefront porch of the sync pulse (see FIG. 5). The complex numbers C and Smay also be computed as above using samples taken on the back porch ofthe sync pulse and averaged with the front porch computations to obtaineven better estimates.

[0045] The phase angle of P corresponds to the phase error, i.e.,difference from the real (I) axis of the phase rotated video signal, andthe magnitude of P is the 100% video signal amplitude, which may be usedfor AGC. The residual DC offset due to the constant of integration inthe I signal is given by the real part of C*P′/|P|, where P′ is theconjugate of P, and the residual DC offset due to the constant ofintegration in the Q signal is given by the real part of S*P′/|P|. Theseresidual DC offset estimates may be scaled and provided to an averagingaccumulator forming a feedback loop integrator where they are subtractedfrom the I and Q signals before phase rotation to reduce the residual DCoffset.

[0046] Referring now to FIG. 7, a homodyne receiver 202 is illustratedthat is configured for use in a television receiver that is centered athalf the color subcarrier frequency above the video carrier inaccordance with embodiments of the present invention. The homodynereceiver 202 comprises a television bandpass filter 204, which may becoupled to an antenna for receiving a television signal. The output ofthe bandpass filter 204 is coupled to a LNA 206. The output of the LNA206 is coupled to the inputs of two mixer circuits 208 and 210. A QVCO212 generates an output signal at half the color subcarrier frequencyabove the video carrier. The output signal from the QVCO 212 is providedas an input signal to both of the mixer circuits 208 and 210. The outputof the mixer circuit 208 is coupled to a low pass filter 214, adifferentiator circuit 216, and an A/D convertor 218 connected in seriesas shown in FIG. 7. Similarly, the output of the mixer circuit 210 iscoupled to a low pass filter 222, a differentiator circuit 224, and anA/D convertor 226. The output of the A/D convertor 218 is coupled to anadder circuit 228 and an accumulator circuit 232, which are connected inseries. Similarly, the output of the A/D convertor 226 is coupled to anadder circuit 234 and an accumulator circuit 236, which are connected inseries. The outputs of the accumulator circuits 232 and 236 are coupledto a rotation circuit 238. A feedback loop is formed by a DSP circuit242, which couples the outputs from the rotation circuit 238 to theinputs of the adder circuits 228 and 234. Exemplary operations of thehomodyne receiver 202, in accordance with embodiments of the presentinvention, will now be described.

[0047] The bandpass filter 204 is configured to receive incomingtelevision signals and to reject other signals outside of this frequencyband. The incoming television signal is amplified by the LNA 206 andprovided to the mixer circuits 208 and 210. The mixer circuit 208multiplies the received television signal by the cosine wave signaloutput from the QVCO 212 to generate the I signal component. Similarly,the mixer circuit 210 multiplies the received television signal by thesine wave signal output from the QVCO 212 to generate the Q signalcomponent. The I and Q signal components are filtered by the low passfilters 214 and 222, respectively, to remove those componentscorresponding to the sum of the incoming video signal frequency and thehalf color subcarrier frequency above the video carrier. The low passfilters 214 and 222 may also include a notch filter for separating theaudio signal from the incoming television signal for processing by asound processing circuit 244.

[0048] The homodyne DC offset may be reduced by differentiating the Iand Q signals using the differentiator circuits 216 and 224, convertingthe differentiated signals to digital samples using the A/D convertors218 and 226, and then integrating the digital samples by digitalaccumulation using the adder circuits 228 and 234 and the accumulatorcircuits 232 and 236.

[0049] The large homodyne offset may, thus, be replaced with anarbitrary constant of integration. This constant may be determined usingclassic differential equation principles by applying boundaryconstraints. The boundary constraints are that the recovered line orframe sync pulses conform to a waveform known a-priori. In particular,the half color subcarrier components occurring during the sync pulsesshould be absent. Detection of non-zero half color subcarriercomponents, as described above with reference to FIG. 6, allow thecomplex amounts C*P′/|P| and S*P′/|P| to be determined and fed back tothe adder circuits 228 and 234 to correct the offset. Feeding backdetermined values to the adder circuits 228 and 234 prior to processingby the accumulator circuits 232 and 236 effectively places theintegration function in the offset correction loop so that a first-orderservo system for annulling the offset is created. Such first ordersystems settle to a steady state having zero error if the parameterbeing controlled is static. Thus, the residual homodyne offset due tothe constant of reintegration may be controlled to zero by the homodynereceiver of FIG. 7 in accordance with embodiments of the presentinvention.

[0050] After the accumulators have reintegrated the signal, the halfcolor subcarrier frequency offset is removed by the rotation circuit238. The rotation circuit 238 provides a derotation function byperforming complex multiplication to rotate the phase of the I and Qvalues successively by the cyclic sequence 0, −45, −90, −135, −180,−225, −270, −315, . . . degrees. After derotation, the sync pulsewaveforms are restored except that they may appear in the I stream, theQ stream, or at a phase in between the real and imaginary axes. The DSPcircuit 242 provides a phase correction function to correct this phaseerror and drift in the output signal from the rotation circuit 238 sothat the sync pulse and the video signal appear in the I waveform. Thephase correction function of the DSP circuit 242 is effectively a phaselocked loop. The estimated frequency error component can be used tocontrol the local oscillator QVCO 212 as AFC. In other embodiments, theQVCO 212 may be controlled by a digital frequency synthesizer using anaccurate crystal reference. The amplitude of the sync pulse may be usedfor AGC, which adjusts analog gain or digital scaling such that the syncpulse amplitude reaches a desired level that is representative of the100% video level. The DSP circuit 242 may also determine the time ofoccurrence of the sync pulses using known sync-separation techniquesthat may involve using a line-scan-rate phase locked loop to lock on tothe periodicity of the sync pulse peaks. This phase locked loop mayoperate on the value of |P| so that it is operable before the angle of Pis corrected by the phase correction loop of the DSP 242. Once thecomposite video signal is obtained, it may be processed digitally toimplement numerically the color subcarrier demodulation and colordemultiplexing to RGB signals normally performed by analog circuitry.

[0051] In the NTSC standard, the frame frequency is just less than the60 Hz line frequency and one complete video field is interlaced over twoframes, which makes the field frequency 29.97 Hz. The number of linesper field is 525, which makes the line scan frequency 525*29.97 or15734.2637 Hz. The color subcarrier is deliberately set midway betweentwo integer multiples of the line scan frequency and is 227.5 times theline scan frequency, i.e., 3579545 Hz. The purpose of this is so thatinterference from the color signal to the luminance signal is offset byhalf a cycle between adjacent lines, reducing its visual impact. In theabove description of homodyne receiver embodiments with reference toFIG. 7, the I and Q signals may be sampled at 4 times the colorsubcarrier frequency, such that sampling is coherent with the colorsubcarrier. The position of the samples with respect to a fixed feature,such as the sync pulse, shifts by half a color subcarrier cycle betweenadjacent lines, which is nevertheless an integer number of 227.5*4=910samples per line. Similar relationships exist with other televisionstandards, such as PAL and SECAM.

[0052] Referring now to FIG. 8, a multi-channel, diversity homodynereceiver 302 is illustrated that is configured for use in a televisionreceiver that is centered at half the color subcarrier frequency abovethe video carrier in accordance with embodiments of the presentinvention. Although only two channels are illustrated in FIG. 8, it willbe understood that the principles and concepts may be applied toadditional channels in accordance with embodiments of the presentinvention. Signals on each channel are processed by homodynedownconvertor circuitry, differentiated, and converted to digitalsamples as discussed above with reference to FIG. 7. The diversityhomodyne receiver 302 may be viewed as having a real-time processingportion and a non-real-time processing portion. The real-time processingportion comprises an adder circuit 304, which receives digital I signalsamples associated with a first channel, an adder circuit 306, whichreceives digital Q signal samples associated with the first channel, anadder circuit 308, which receives digital I signal samples associatedwith a second channel, and an adder circuit 312, which receives digitalQ signal samples associated with the second channel. A pair ofaccumulator circuits 314 and 316 is coupled to the outputs of the addercircuits 304 and 306, respectively. Similarly, a pair of accumulatorcircuits 318 and 322 is coupled to the outputs of the adder circuits 308and 312, respectively. A first rotation circuit 324 couples the outputsof the accumulator circuits 314 and 316 to a first phase correctioncircuit 326. A second rotation circuit 328 couples the outputs of theaccumulator circuits 318 and 322 to a second phase correction circuit332.

[0053] The outputs from the phase correction circuits 326 and 328 arecoupled to a memory buffer 334, which comprises part of thenon-real-time processing portion of the diversity homodyne receiver 302.The non-real-time processing portion of the diversity homodyne receiver302 further comprises a line synchronization circuit 336, a complexdigital filter circuit 338, a channel estimation circuit 342, a phaseand frequency error circuit 344, and a homodyne offset estimator circuit346, which are configured as shown in FIG. 8. Exemplary operations ofthe diversity homodyne receiver 302, in accordance with embodiments ofthe present invention, will now be described.

[0054] Operations of the real-time processing portion of the diversityhomodyne receiver 302 are similar to that described above with referenceto FIG. 7 for the homodyne receiver 202. The phase corrected outputsignals associated with the two channels maybe denoted by S1=I1+jQ1 andS2=I2+jQ2. These two output signals may be stored in the memory buffer334. The memory buffer 334 may, for example, be implemented as a cyclicbuffer, such as a one or two line memory, or as a larger memory unit.The output signals S1 and S2 may be processed in non-real-time byreading values from the memory buffer 334, operating on these values,and rewriting the values to the memory buffer 334 so that the values maybe processed multiple times to extract different properties.Time-reversed processing may be particularly useful to compensate forghosting, and the use of diversity may permit relatively long ghostdelays to be compensated for.

[0055] As shown in FIG. 8, the signals S1 and S2 may be filtered by thecomplex digital filter circuit 338 using both finite impulse response(FIR) filters and infinite impulse response (IIR) filters. The complexvideo sample data S1 from channel 1 is FIR filtered using a filtermatched to channel 1, which is denoted by C1 ^(#) and in the time domainis a time-reversed conjugate filter formed from the coefficients of thez-polynomial C1 that describes the first multipath channel. The complexvideo sample data S2 from channel 2 is FIR filtered using a filtermatched to channel 2, which is denoted by C2 ^(#) and in the time domainis a time-reversed conjugate filter formed from the coefficients of thez-polynomial C2 that describes the second multipath channel. The sum ofthe signals from the FIR filters is then IIR filtered using an IIRfilter having the following transfer function: 1/(C1 ^(#)C1+C2 ^(#)C2).

[0056] The denominator of the IIR filter transfer function has rootsthat occur in conjugate-reciprocal pairs. Thus, half of the roots have amagnitude less than one and may be applied through forward timeprocessing. The other half of the roots have magnitudes greater thanone, i.e., are non-causal, and may be implemented using an IIR filterthat is formed from the reciprocal roots applied to the video sampledata taken in time-reversed order. The IIR filters have, in principle,infinite memory, but these filters may be used to process one line ofvideo samples at a time as follows:

[0057] (i) initialize the time-reverse IIR filter memory by jamming inthe expected sync pulse waveform, which is known a-priori;

[0058] (ii) run the IIR filter initialized in (i) over the video samplestaken in time-reversed order, starting with the sample immediatelypreceding the sync pulse. Continue processing through all samples of thepreceding sync pulse;

[0059] (iii) initialize the forward-time IIR filter memory by jamming inthe expected sync pulse waveform of the preceding sync pulse knowna-priori and as many already-processed video samples from the precedingline as may be desired;

[0060] (iv) run the IIR filter initialized in (iii) from the first videosample after the sync pulse referred to in (ii) and continue processingthrough to the end of the next sync pulse.

[0061] After the above four operations, the “preceding sync pulse”mentioned in (ii) and (iii) will have been processed in both forward andreverse time order, and its waveform may be processed for determiningother features.

[0062] To implement the IIR filters, the roots of the polynomial C1^(#)C1+C2 ^(#)C2 are calculated. The roots may also recalculatedwhenever C1 or C2 changes. The process of updating the estimates of C1and C2 and recalculating the roots may be simplified by avoiding anentire recalculation and instead performing a root update procedure totake into account small changes in C1 and C2. A method of determiningcomplex roots of polynomials with complex coefficients is described inU.S. patent application Ser. No. 09/915,896, filed Jul. 26, 2001, thedisclosure of which is hereby incorporated herein by reference. Themethod is based on determining the complex amount by which root(i) is inerror, given current approximations for the roots from Equation 2 below:$\begin{matrix}\frac{P( {{root}(i)} )}{\prod\limits_{k \neq i}^{\quad}\quad \lbrack {{{root}(k)} - {{root}(i)}} \rbrack} & {{EQ}.\quad 2}\end{matrix}$

[0063] where P is the polynomial to be factorized. When the roots are tobe found for the first time given a new set of coefficients forpolynomial P, this Equation 2 is computed iteratively starting withinitial values for the roots and then incrementing the index i from 1through 2N, where N is the maximum multipath channel delay in videosample periods.

[0064] If the coefficients change to a completely different set, thenthe iterative process of root finding begins anew. If, however, thecoefficients change slightly, as when channel estimates are updated fromprevious values to new values based on more received signal samples, theiterative process need not necessarily begin anew. Rather, with theassumption that the new roots for the updated polynomial will only havemoved slightly from the old root positions, a single iteration of theabove formula may suffice to update the roots. It may, therefore, bepossible to perform root tracking as well as channel tracking usingEquation 2. In television receivers, when a new line of video samples isreceived, the operations described above may be performed to equalizeone or more lines of video signal data to update the channelcoefficients from a set C1, C2 to a new set C1+dC1, C2+dC2. The newcoefficients are used to reform the polynomial P, which will in turnhave its coefficient set updated to the set P+dP. Calculating Equation 2one time for each value of i using the updated coefficients P+dPdetermines the error in the roots, i.e., the amount by which a root hasto be changed to refine its value so that it is a root of the updatedpolynomial coefficients. Because the roots of P occur in conjugatereciprocal pairs, Equation 2 need only be calculated N times, each timeupdating a root and its conjugate reciprocal partner. For ghost delaysequal to a significant fraction (e.g., 30%) of a line period (e.g., 273samples), a relatively fast processor may be needed. The frequency ofthe root update operations is linked to the speed at which multipathcoefficients change. Thus, root update operations need not be performedevery line scan period. It may be sufficient to perform root updateoperations once per frame period or less for static televisionreceivers, while once per ten line scan periods may be sufficient formobile television reception. One compromise is to execute Equation 2 toupdate only one root per line scan period, such that all of the possible273 root pairs are updated each frame period.

[0065] Separate from the root tracking operations described above, theFIR and IIR filters may be implemented with a length of 273 and operateat the video sample rate, which may be processing intensive. Suchfilters, however, may be implemented in hardware with their coefficientsprogrammable in accordance with embodiments of the present invention.

[0066] Returning to FIG. 8, The magnitude squared of the sync pulsesamples is given by E²+F², where E=S1 C1 ^(#)/(C1 ^(#)C1+C2 ^(#)C2) andF=S2 C2 ^(#)/(C1 ^(#)C1+C2 ^(#)C2). This magnitude may then be used bythe line synchronization circuit 336 to adjust line synchronization.Initially, before any estimate of line sync is available, a search ofthe processed samples may be made for peak video values indicative ofthe likely position of sync pulses. Thereafter, the estimate of the syncpulse position in the cyclic buffer memory may be refined by estimatingwhere rising and falling edges lie and adjusting the mean sync pulseposition estimate to coincide with the rising and falling edges.

[0067] Having determined the sync pulse position, the channel estimationcircuit updates the estimates of the channel polynomials C1 and C2 bycorrelating the unprocessed channel signals S1 and S2 with the expectedsync pulse waveform. In accordance with some embodiments of the presentinvention, the expected sync pulse waveform may include the colorsubcarrier reference burst. In other embodiments, instead of correlatinga channel signal with a sync pulse waveform, a filtered channel waveformmay be correlated with a similarly filtered sync pulse waveform, using afilter, such as a differentiator, to enhance edges, thereby givingimproved time resolution accuracy. More specifically, the filter mayhave the effect of making the spectrum of the filtered sync pulsewaveform as close to white as possible. Such a filter may be precomputedalong with the filtered sync pulse waveform used for correlation.

[0068] The processed, diversity combined sync pulse samples may also beused by the phase and frequency error circuit 344 to correct the phaseof the video waveform so that the video signal lies in the real plane.The phase angle of the samples lying at the sync pulse peak may bedetermined and fed to a second order digital phase locked loop thatestimates both phase and frequency error from rate of change of phase byconventional techniques. A common phase rotation value is sent to thephase correction units of all channels, which may lie in the real-timesignal-processing portion of the diversity homodyne receiver 302 (e.g.,first and second phase correction circuits 326 and 332). The phasecorrection may be computed by applying the frequency error to a modulo2π accumulator so that phase corrections are updated continuouslybetween instants when the frequency error is updated.

[0069] The homodyne offset, which is now, as discussed above, replacedby an arbitrary constant of integration, is estimated by the homodyneoffset estimator circuit 346. The offset may be estimated by correlatingthe signals S1 and S2 with the half color subcarrier frequency signal,as described above with reference to FIG. 7. The offset values dI1, dQ1,dI2, and dQ2 may then be applied by the adder circuits 304, 306, 308,and 312, respectively, to form a first-order offset nulling loop.

[0070] After the operations described above, the real part E of theprocessed video samples may be used to perform chrominance signalextraction, demodulation, and combination with the luminance signal toform RGB signals.

[0071] Analog television transmission conforming to the NTSC, PAL,and/or SECAM standards may be supplemented by digital televisionbroadcasting. Digital video broadcasting may provide higher definition(e.g., HDTV) and better quality. The use of equalizers to decode digitalsignals in the presence of multipath propagation distortion is generallymore widely understood and practiced than analog television ghostcompensation described above. The principles and concepts describedherein, however, may be used as equalization techniques for digitaltelevision signals. Moreover, diversity reception may also be beneficialin reducing errors in digital signals. For example, the diversityhomodyne receiver of FIG. 8 may be used for digital television receptionby modifying the line synchronization circuit to operate as a digitaldecoder for decoding binary data from the signal using, for example,error correction decoding. The analogous function to linesynchronization may be acquiring frame synchronization with the digitalsignal by detecting the presence of a known symbol pattern in thedigital stream. A known symbol pattern inserted periodically at thetransmitter may also be used for channel estimation in place of usingthe sync pulse for channel estimation in an analog televisionenvironment. Such known symbol patterns may be called sync words in theart of digital signal equalization as practiced in digital cellulartelephone systems, for example. Thus, the block diagrams of FIGS. 4, 7,and 8 may be used for digital and analog television reception byincorporating digital process algorithms along with analog processingalgorithms in the same hardware units or chips.

[0072] Embodiments of television receivers have been described hereinthat may operate based on homodyne reception and digital signalprocessing and may be implemented using a single radio frequency chipand a single digital signal-processing chip. Moreover, these receiverembodiments may be readily designed to decode analog television standardsignals and digital or high definition television standard signals byincorporating appropriate modes within the digital chip and controllingit to decode a selected standard. Thus, a multi-mode analog/digital ormulti-standard television receiver may be constructed usingsubstantially the same components for each mode or standard. Embodimentsof television receivers, in accordance with the present invention, mayalso accommodate diversity reception and reduce ghosting. Diversityequalization techniques described above may also be used to improvepicture stability during mobile reception.

[0073] In concluding the detailed description, it should be noted thatmany variations and modifications can be made to the preferredembodiments without substantially departing from the principles of thepresent invention. All such variations and modifications are intended tobe included herein within the scope of the present invention, as setforth in the following claims.

I claim:
 1. A multi-mode television picture receiving apparatus,comprising: a radio frequency (RF) circuit that is configured togenerate a complex baseband signal responsive to a television picturesignal received from a television transmitter; an analog-to-digital(A/D) converter that generates digital samples responsive to the complexbaseband signal; and a digital signal processing (DSP) circuit that isconfigured to generate decoded color video signals responsive to thedigital samples, the DSP circuit being switchable between a firstdecoding mode for an analog television standard and a second decodingmode for a digital television standard.
 2. The apparatus of claim 1,wherein the RF circuit is a direct conversion (homodyne) receivercircuit.
 3. The apparatus of claim 1, wherein the DSP circuit isconfigured to remove a homodyne direct current (DC) offset from thedigital samples.
 4. The apparatus of claim 1, wherein the RF circuitcomprises two receiver channels for receiving the television picturesignal from two different antennas and is further configured to generatea first complex baseband signal responsive to the television picturesignal received through a first one of the antennas and to generate asecond complex baseband signal responsive to the television picturesignal received through a second one of the antennas, wherein the A/Dconverter is further configured to generate first digital samplesresponsive to the first complex baseband signal and second digitalsamples responsive to the second complex baseband signal, and whereinthe DSP circuit is further configured to combine the first and thesecond digital samples to generate the decoded color video signals. 5.The apparatus of claim 4, wherein the DSP circuit is further configuredto filter the first and the second digital samples to reduce ghosting inthe decoded color video signals.
 6. The apparatus of claim 4, whereinthe DSP circuit comprises an equalizer circuit that is configured toprocess the first and the second digital samples to reduce ghosting inthe decoded color video signals.
 7. A television picture receivingapparatus, comprising: a direct conversion radio frequency (RF) circuitthat comprises a quadrature converter circuit, the quadrature convertercircuit being configured to generate a complex baseband signalresponsive to a television picture signal received from a televisiontransmitter; an offset canceller circuit that is configured to attenuatea direct current (DC) offset in the complex baseband signal; ananalog-to-digital (A/D) converter that is configured to generatenumerical samples responsive to the complex baseband signal; and adigital signal processing (DSP) circuit that is configured to generatedecoded color video signals responsive to the numerical samples.
 8. Theapparatus of claim 7, wherein the DSP circuit is switchable between afirst decoding mode for a first television standard and a seconddecoding mode for a second television standard.
 9. The apparatus ofclaim 8, wherein the first television standard is an analog transmissionstandard and the second television standard is a digital televisiontransmission standard.
 10. The apparatus of claim 9, wherein the analogtransmission standard is NTSC.
 11. The apparatus of claim 9, wherein thedigital television transmission standard uses 8-level vestigial sidebandamplitude modulation.
 12. The apparatus of claim 7, wherein thedirection conversion RF circuit comprises two receiver channels forreceiving the television picture signal from two different antennas,wherein the quadrature convertor circuit is further configured togenerate a first complex baseband signal responsive to the televisionpicture signal received through a first one of the antennas and togenerate a second complex baseband signal responsive to the televisionpicture signal received through a second one of the antennas, whereinthe A/D converter is further configured to generate first numericalsamples responsive to the first complex baseband signal and secondnumerical samples responsive to the second complex baseband signal, andwherein the DSP circuit is further configured to combine the first andthe second numerical samples to generate the decoded color videosignals.
 13. The apparatus of claim 12, wherein the DSP circuitcomprises an equalizer circuit that is configured to process the firstand the second numerical samples to reduce ghosting in the decoded colorvideo signals.
 14. The apparatus of claim 13, wherein the equalizercircuit comprises: a memory having ones of the first and the secondnumerical samples associated with at least one horizontal line scanperiod stored therein; and wherein the equalizer circuit is configuredto process the ones of the first and the second numerical samples intime-reversed order.
 15. The apparatus of claim 7, wherein thetelevision picture signal comprises a video carrier frequency signal anda color subcarrier frequency signal, and wherein the quadratureconvertor circuit comprises a local oscillator that generates an outputsignal at a frequency that is displaced from the video carrier frequencyby half of the color subcarrier frequency.
 16. A method of operating atelevision receiver, comprising: receiving a television signal;downconverting the television signal to generate a complex basebandsignal; differentiating the complex baseband signal; and integrating thedifferentiated complex baseband signal to generate an integrated outputsignal.
 17. The method of claim 16, further comprising: sampling thedifferentiated complex baseband signals to generate digital samplesthereof; and wherein integrating the differentiated complex basebandsignal comprises integrating the digital samples.
 18. The method ofclaim 16, wherein downconverting the television signal to generate thecomplex baseband signal comprises: multiplying the television signal bya local oscillator signal having a frequency that is different from acarrier frequency of the television signal.
 19. The method of claim 18,wherein the television signal comprises a color subcarrier frequency andthe local oscillator signal is equal to the carrier frequency of thetelevision signal plus half of the color subcarrier frequency.
 20. Themethod of claim 19, further comprising: rotating the integrated outputsignal in the complex domain to remove a frequency offset correspondingto half of the color subcarrier frequency.
 21. The method of claim 16,wherein the television signal comprises a periodic known informationfield, the method further comprising: adjusting a phase of theintegrated output signal so that the known information field is on thereal axis.
 22. The method of claim 16, wherein the television signalcomprises a periodic known information field, the method furthercomprising: correlating the integrated output signal with the knowninformation field to determine a residual offset therein associated witha constant used in integrating the differentiated complex basebandsignal; and subtracting the residual offset from the constant used inintegrating the differentiated complex baseband signal.
 23. A method ofoperating a television receiver, comprising: receiving a firsttelevision signal on a first channel, the first television signalcomprising a first known information field; receiving a secondtelevision signal on a second channel, the second television signalcomprising a second known information field; downconverting the firsttelevision signal to generate a first complex baseband signal;downconverting the second television signal to generate a second complexbaseband signal; storing samples of the first and second complexbaseband signals in a memory; and filtering the stored samples of thefirst and second complex baseband signals by processing the storedsamples in forward and reverse time order.
 24. The method of claim 23,wherein filtering the stored samples comprises: filtering the storedsamples of the first complex baseband signal using a first finiteimpulse response (FIR) filter having a transfer function matched to thefirst channel; and filtering the stored samples of the second complexbaseband signal using a second finite impulse response (FIR) filterhaving a transfer function matched to the second channel.
 25. The methodof claim 24, wherein filtering the stored samples further comprises:summing outputs of the first and second FIR filters; filtering thesummed outputs of the first and second FIR filters using an infiniteimpulse response (IIR) filter having a transfer function given by 1/(C1^(#)C1+C2 ^(#)C2), where C1 corresponds to the coefficients of az-polynomial describing the first channel, C1 ^(#) corresponds to thecoefficients of the first FIR filter, C2 corresponds to the coefficientsof a z-polynomial describing the second channel, and C2 ^(#) correspondsto the coefficients of the second FIR filter.
 26. The method of claim23, further comprising: determining positions of the first and secondknown information fields based on the filtered stored samples.
 27. Themethod of claim 23, further comprising: estimating transfer functions ofthe first and second channels based on the filtered stored samples bycorrelating the first and second complex baseband signals with the firstand second known information fields.
 28. The method of claim 23, furthercomprising: differentiating the first and second complex basebandsignals; and integrating the differentiated complex baseband signals.29. A television receiver, comprising: a downconvertor circuit thatgenerates a complex baseband signal responsive to a television signal; adifferentiator circuit that generates a differentiated complex basebandsignal responsive to the complex baseband signal; and an integratorcircuit that generates an integrated output signal responsive to thedifferentiated complex base band signal.
 30. The television receiver ofclaim 29, wherein the integrator circuit comprises an adder circuit andan accumulator circuit connected in series.
 31. The television receiverof claim 30, further comprising an analog-to-digital (A/D) convertorthat couples the differentiator circuit to the adder circuit.
 32. Thetelevision receiver of claim 30, further comprising: a processor circuitthat is configured to correlate the integrated output signal with aknown information field to determine a residual offset therein, theadder circuit being responsive to the residual offset.
 33. Thetelevision receiver of claim 29, further comprising: a rotation circuitthat rotates the integrated output signal in the complex domain toadjust a frequency offset in the integrated output signal.
 34. Atelevision receiver, comprising: a first downconvertor circuit thatgenerates a first complex baseband signal responsive to a firsttelevision signal received on a first channel, the first televisionsignal comprising a first known information field; a seconddownconvertor circuit that generates a second complex baseband signalresponsive to a second television signal received on a second channel,the second information signal comprising a second known informationfield; a memory that has samples of the first and second complexbaseband signals stored therein; and a filter circuit that filters thestored samples of first and second complex baseband signals in forwardand reverse time order.
 35. The television receiver of claim 34, whereinthe filter circuit comprises: a first finite impulse response (FIR)filter that has a transfer function matched to the first channel andgenerates a first output signal responsive to the stored samples of thefirst complex baseband signal; a second finite impulse response (FIR)filter that has a transfer function matched to the second channel andgenerates a second output signal responsive to the stored samples of thesecond complex baseband signal; and an infinite impulse response (IIR)filter that has a transfer function given by 1/(C1 ^(#)C1+C2 ^(#)C2),where C1 ^(#) corresponds to the coefficients of a z-polynomialdescribing the first channel, C1 ^(#) corresponds to the coefficients ofthe first FIR filter, C2 corresponds to the coefficients of az-polynomial describing the second channel, and C2 ^(#) corresponds tothe coefficients of the second FIR filter, and generates a third outputsignal that is responsive to a sum of the first and second outputsignals of the first and second FIR filters, respectively.
 36. Thetelevision receiver of claim 34, further comprising: a synchronizationcircuit that determines positions of the first and second knowninformation fields responsive to the filtered stored samples.
 37. Thetelevision receiver of claim 34, further comprising: a channelestimation circuit that estimates transfer functions of the first andsecond channels responsive to the filtered stored samples by correlatingthe first and second complex baseband signals with the first and secondknown information fields, the filter circuit being responsive to theestimated transfer functions of the first and second channels.